1. Field of the Invention
The present invention relates to circuits for power conversion applications. In particular, the present invention relates to a single-stage input-current-shaping converter for universal line-range applications.
2. Description of the Related Art
The harmonic content of a current drawn by an electronic equipment from the AC mains is regulated by a number of standards. To comply with these standards, input-current shaping (ICS) of off-line power supplies is necessary. At present, various passive and active ICS techniques are used. While the passive techniques are preferred in many cost-sensitive applications, active ICS techniques are used in most applications because of their superior performance.
The most commonly used active approach that meets both high power and high quality requirements is the "two-stage" approach. In this approach, a non-isolated boost-like converter, which is controlled so that the rectified line current follows the rectified line voltage, is used as the input stage that creates an intermediate DC bus with a relatively large ripple at the second-harmonic frequency of the line. This ICS stage is then followed by a DC/DC converter which provides isolation and high-bandwidth voltage regulation. The DC/DC converter attenuates the second-harmonic ripple to an acceptable level. At a high-power level, the ICS stage is operated in the continuous-conduction mode (CCM), while at a lower power level, a discontinuous-conduction mode (DCM) is commonly used. Compared to CCM, DCM is simpler to control.
To reduce component count and to improve performance, a number of "single-stage" ICS techniques are introduced recently. Under a single-stage approach, input-current shaping, isolation, and high-bandwidth control are performed in a single step (i.e., without creating an intermediate DC bus). Generally, a single-stage ICS converter uses an internal storage capacitor to buffer between an instantaneously varying input power and a constant output power.
Among single-stage input-current-shaping (S.sup.2 ICS) circuits are a number of circuits described in the following publications: (a) "A Switching Power Supply of 99% Power Factor by the Dither Rectifier," by I. Takahasi et al., IEEE International Telecommunications Energy Conf. (INTELEC) Proc., pp. 714-719, November 1991; (b) "Integrated High-Quality Rectifier-Regulators," by M. Madigan et al., IEEE Power Electronics Specialists Conf. (PESC) Record, pp. 1043-1051, June 1992; (c) U.S. Pat. No. 5,301,095, entitled "High Power Factor AC/DC Converter," to S. Teramoto et al., filed on Sep. 28, 1992, and issued on Apr. 5, 1994; (d) "The Suppressing Harmonic Currents, MS (Magnetic Switch) Power Supply," by H. Watanabe et al., IEEE International Telecommunication Energy Conf. (INTELEC) Proc., pp. 783-790, October 1995; (e) U.S. Pat. No. 5,600,546, entitled "Input Harmonic Current Corrected AC-to-DC Converter with Multiple Coupled Primary Windings," to F. M. S. Ho et al., filed on Oct. 16, 1996, and issued on Feb. 4, 1997; (f) U.S. Pat. No. 5,652,700, entitled "Low Cost AC-to-DC Converter Having Input Current with Reduced Harmonics," to F. S. Tsai et al, filed on Jan. 19, 1996, and issued on Jul. 29, 1997; (g) "A High Efficient Single Stage Single Switch High Power Factor AC/DC Converter with Universal Input," by J. Qian et al., IEEE Applied Power Electronics Conference (APEC) Proc., pp. 281-287, February 1997; (h) U.S. Pat. No. 5,757,626, entitled "Single-Stage, Single-Switch Isolated Power-Supply Technique with Input-Current Shaping and Fast Output-Voltage Regulation," to M. M. Jovanovic at al, filed on Oct. 4, 1996, and issued on May 26, 1998; and (i) U.S. Pat. No. 5,790,389, entitled "Consolidated Soft-Switching AC/DC Converters," to G. Hua, filed on May 31, 1996, and issued on Aug. 4, 1998.
For many applications, single-switch S.sup.2 ICS converters are particularly attractive because they can be implemented with only one semiconductor switch and a simple control. In the single-switch S.sup.2 ICS circuits described in the references above, the single-switch S.sup.2 ICS converters integrate a boost-converter front-end with a forward-converter or a flyback-converter DC/DC stage. For example, FIGS. 1-3 show single-switch S.sup.2 ICS converters described in the above-referenced publications of Teramoto et al , Watanabe at al., and Tsai et al., respectively.
As shown in FIGS. 1-3, the front-ends of converters 100, 200, and 300 each include a full-bridge rectifier 101 connected to a boost converter including boost inductor 102, boost rectifier 103, and storage (bulk) capacitor 104. The output portions of converters 100, 200, and 300 are conventional DC/DC, single-switch forward or flyback converters. In converter 100 of FIG. 1, boost inductor 102 is energized through capacitor 110 when switch 105 is closed. In converters 200 and 300 of FIGS. 2 and 3, boost inductors 102 are energized through windings 111 of transformers 206 and 306, respectively.
In converter 100 of FIG. 1, boost inductor 102 operates in CCM, while in each of converters 200 and 300 of FIGS. 2 and 3, boost inductor 102 operates in DCM. The CCM operation offers a slightly higher efficiency over the DCM operation. However, the DCM operation provides a lower total harmonic distortion (THD) of the line current over the CCM operation.
As mentioned above, in an ICS application, the rectified line voltage contains a large ripple. This ripple propagates through the power stage, causing an increased output-voltage ripple at the rectified-line frequency (i.e., the second-harmonic frequency of the line). To eliminate the rectified-line-voltage component of the output-voltage ripple, the output-voltage feedback loop is designed with a bandwidth which is wide enough to attenuate the ripple to a desired value. The desired bandwidth, regulation accuracy, and control-loop stability are set by a proper selection of the voltage-loop compensation.
Although it has been demonstrated that the S.sup.2 ICS converters described in the above-referenced publications can achieve the desired performance in a variety of applications, power supplies based on these approaches have significant difficulties meeting performance expectations in universal-line (e.g., 90-270 V AC) applications with a hold-up time requirement. For example, most of today's desktop computers and computer peripherals require power supplies that are capable of operating in the 90-270 V AC range and can provide a hold-up time of at least 10 ms. Generally, the hold-up time is the time during which a power supply must maintain its output voltage within a specified range after a drop-out of the line voltage. The hold-up time is used to orderly terminate the operation of a computer or to switch over to an uninterruptible-power-supply (UPS) operation after a line failure. The required energy to support the output voltage during the hold-up time is obtained from a properly-sized storage capacitor, such as capacitor 104. Since the energy stored in a capacitor is proportional to its capacitance and the square of the voltage across its terminals, the required capacitance increases with an increase of the hold-up time. Also, the same hold up time can be achieved by a smaller capacitance, if the voltage across the storage capacitor's terminals is higher.
Since voltage V.sub.C across storage capacitor 104 varies with both the line voltage and the load current, converters 100, 200, and 300 in FIGS. 1-3 have difficulty in satisfying the wide line range and long hold-up time requirements. Specifically, in converters 100, 200, and 300, only output voltage V.sub.o across terminals 107 and 108 is regulated. Voltage V.sub.C across storage capacitor 104 follows the root-mean-square (rms) variations of the line voltage. Therefore, since the universal-line range is 3:1 (90-270 V AC), voltage V.sub.C also varies over a range 3:1. In addition, because storage capacitor 104 is the output filter of the boost converter, voltage V.sub.C must be higher than the peak of the line voltage. As a result, for a line voltage of 270 V AC, voltage V.sub.C is at least in the 380-390 V DC range. Moreover, storage capacitor voltage V.sub.C increases as the load current decreases. In most applications, voltage V.sub.C can be kept in the 410-420 V DC range. Thus, capacitor 104 can be implemented by a 450 V electrolytic capacitor. Since the capacitance of capacitor 104 is determined from the hold-up time requirement at the minimum line voltage (worst case), converters 100, 200, and 300 of FIGS. 1-3 each require a relatively bulky and expensive storage capacitor. Further, because voltage V.sub.C, which represents an input voltage to the DC/DC output stage, varies over a wide range, conversion efficiency of the output stage is reduced. In contrast, under the two-stage approach, in which the storage capacitor voltage is independently regulated at approximately 380 V DC, a much smaller and, therefore, cheaper electrolytic capacitor rated at 450 V, or even 400 V, is adequate. In addition, in a two-stage approach, due to the regulated storage capacitor voltage V.sub.C, the efficiency of the DC/DC output stage can be made higher than the efficiency of the single-stage approach in the converters of FIGS. 1-3. Relative to the two-stage approach, because of the size and the cost of the power supply necessary, the relatively large capacitor (i.e., storage capacitor 104) required for a S.sup.2 ICS converter is a significant drawback.
The performance of a conventional, universal-line-range power supply without ICS can be improved by a voltage-doubler rectifier, such as converter 400 shown in FIG. 4. In FIG. 4, converter 400 includes a voltage-doubler rectifier, which operates as a conventional full-bridge rectifier when range-select switch 401 is open, and as a voltage doubler when range-select switch 401 is closed. Generally, range-select switch 401 can be a mechanical or an electronic switch. When converter 400 is connected to a power line with a nominal line voltage of 220/240 V AC (European line), switch 401 is open. Conversely, when converter 400 operates from a power line with the nominal voltage of 100/120 V AC, switch 401 is closed. With range-select switch 401 in FIG. 4 open (i.e., operating from a 220/240 V AC power line), the front-end portion of converter 400 works as a conventional rectifier, and thus the output voltage V.sub.C across the series-connected capacitors 402 and 403 is approximately equal to the peak of the line voltage.
In the voltage-doubler mode (i.e., when switch 401 is closed and converter 400 operates from the 100/120 V AC power line), capacitor 402 is charged to the peak of the line voltage through rectifier 109a during a positive half-cycle of the line voltage, and capacitor 403 is charged to the peak of the line voltage through rectifier 109c during a negative half-cycle of the line voltage. In the voltage-doubler mode, rectifiers 109b and 109d do not conduct, being reverse-biased by voltages V.sub.C1 and V.sub.C2 of capacitors 402 and 403, respectively. Because output voltage V.sub.C of the front-end portion of converter 400 is the sum of capacitor voltages V.sub.C1 and V.sub.C2, (i.e., V.sub.C =V.sub.C1 +V.sub.C2), output voltage V.sub.C in the voltage-doubler mode is twice the peak of the line voltage. Therefore, due to voltage-doubling, output voltage V.sub.C of the front-end portion of converter 400 is approximately the same for both 100/120 V AC and 220/240 V AC power lines. Specifically, for the universal-line range 90-270 V AC, output voltage V.sub.C f the front-end of converter 400 varies from approximately 180 V AC to 270 V AC. Since this resulting voltage range is much narrower than the corresponding voltage range of the conventional wide-range full-bridge rectifier, the conversion efficiency of the DC/DC output portion of converter 400 is improved. In addition, because voltage V.sub.C of converter 400 is at least twice as high as that of the wide-range full-bridge rectifier, the total capacitance (i.e., the sum of the capacitances of capacitors 402 and 403) required for a given hold-up time of converter 400 is approximately one-half of that required in the wide-range full-bridge rectifier. Finally, since each capacitor sees a lower voltage across its terminals, storage capacitors 402 and 403 need only be rated at 250 V DC, or even 200 V DC. Capacitors with a lower voltage rating are usually significantly less expensive than their higher voltage rated counterparts.
Therefore, in a universal-line application with a old-up time requirement, a front-end with voltage-doubler rectifier offers a number of advantages over a conventional, wide-range, full-wave rectifier such as higher conversion efficiency, smaller size, and lower cost.